Chopper-stabilized operational amplifier including integrated circuit with true random voltage output

ABSTRACT

An improved chopper stabilized operational amplifier is disclosed, along with an improved method of timing the switchings of chopper switches in such an amplifier. The disclosure includes an integrated circuit and method for generating a true random voltage signal having a truly random RMS voltage value within a selected range. The true random voltage signal is obtained by amplifying and bandpass filtering random white noise voltages generated by a component of the circuit. The white noise voltages include shot noise voltages generated by bipolar transistors in an input amplifier stage. The random signal generator circuit and method is employed with an oscillator to form a random clock signal generator on the integrated circuit chip. The amount of time between each clocking pulse output by the random clock signal generator truly randomly varies within a selected range of time, and repeats only by random chance. The random clock signal generator is used to control the amount of time between each switching of a chopper switch in a chopper-stabilized operational amplifier, so that the chopping frequency is truly random. The truly random variation of the chopper switching frequency reduces interference due to aliasing of noise typically created by such switches.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of patent application Ser. No.08/810,095, which was filed Mar. 3, 1997, now U.S. Pat. No. 5,926,066.This application is related to three commonly invented and owned U.S.patent applications: application Ser. No. 08/527,400, entitled "DCIsolated Oscillator," which was filed on Sep. 13, 1995, now U.S. Pat.No. 5,600,283 issued on Feb. 4, 1997; application Ser. No. 08/527,401,entitled "Floating Capacitor Differential Integrator," which was alsofiled on Sep. 13, 1995, now U.S. Pat. No. 5,793,242 issued on Aug. 11,1998; and application Ser. No. 08/811,063, entitled "Chopper-StabilizedOperational Amplifier Including Low Noise Chopper Switch," which wasfiled Mar. 3, 1997, now U.S. Pat. No. 5,959,498 issued on Sep. 28, 1999.Each of the foregoing applications is incorporated herein by referencein its entirety.

BACKGROUND

1. Field of the Invention

This invention relates to integrated circuits, and in particular to amethod and circuit for generating a true random, yet controlled, voltagesignal which may be used in clock circuits and chopper-stabilizedoperational amplifiers.

2. Description of Related Art

Attributes of the present invention may be appreciated by consideringthe example of a chopper-stabilized operational amplifier, andconventional approaches to improving the performance of such amplifiers.

Operational amplifiers or "op amps" typically have two input terminals:a noninverting terminal (denoted "+") and an inverting terminal (denoted"-"). When both input terminals are grounded, an ideal operationalamplifier develops a zero output voltage. Under the same conditions, anactual amplifier will have a finite output because of inevitable smallimbalances in the components. The offset voltage of the amplifier equalsthe DC input voltage required to compensate for such imbalances.Unfortunately, such offset voltages "drift" with changes in temperatureand time. This is one of the important sources of error in operationalamplifier circuits, because it is dynamic, and cannot easily ornecessarily be resolved by the initial design of the op amp.

One approach to compensate for voltage drift is to employ achopper-stabilizer circuit. The chopper-stabilizer circuit dynamicallynulls the offset of a primary op amp. There are a variety of ways toimplement a chopper-stabilized operational amplifier. See, e.g., Jonesand Webb, "Chopper-Stabilized Op Amp Combines MOS and Bipolar Elementson One Chip," Electronics, Sep. 27, 1973, pp. 209-13.

FIG. 1 shows a conventional chopper-stabilized operational amplifier 1,with an auxiliary chopper-stabilizer circuit 17. Primary op amp 10 istypically a high-frequency, wideband amplifier. As shown, op amp 10 hasa negative feedback path 16 between its output 18 and its inverting("-") input terminal. Op amp 10 is also provided with a trim port 11, sothat its offset may be nulled. Chopper-stabilizer circuit 17 isconnected between nodes A, B and trim port 11. The offset voltage of opamp 10 is depicted as being a small dc voltage V_(OS), between thenoninverting terminal ("+") and inverting terminal ("-") of op amp 10.

In FIG. 1, the noninverting and inverting terminals of op amp 10 areelectrically connected to chopper-stabilizer circuit 17 via nodes A andB, respectively. At nodes A and B, V_(OS) is converted to a periodicsquare wave by the switching action of switch 12 between nodes A and B.The square wave passes through coupling capacitor C1, and is input toamplifier 13, which amplifies the AC signal. The amplified AC signalthen passes through coupling capacitor C2. An amplified DC voltage isrecovered by the switching action of switch 14 across resistor R1 atnodes D and E, and by a low pass filter comprised of resistor R2 andcapacitor C3. The DC voltage is then input via line 19 into trim port 11of op amp 10 so that the offset voltage of op amp 10 is nulled.

Switches 12 and 14 are synchronized to switch repeatedly between a firststate or position (e.g., nodes A and D), and a second state or position,(e.g., nodes B and E), and vice versa, with an interval of time betweeneach of the switchings that is determined by clock 15. Clock 15 outputsa clock signal which, directly or indirectly, indicates the time atwhich switches 12 and 14 are to switch from their first state to theirsecond state, or vice versa. Clock 15 may be an oscillator, for example,and the clock signal may be, or be formed from, the oscillating outputvoltage states of such an oscillator.

Typically, the amount of time between the switchings of the chopperswitches is constant. That is, the switching frequency is constant. Theswitching frequency is typically a frequency far in excess of thefrequency of the external signal applied to the input(s) of primary opamp 10. This approach reduces interference due to noise generated by theswitches at the switching frequency. This approach is not optimal,however, because of "aliasing," which occurs when harmonics of theswitching noise interfere with the desired signal.

U.S. Pat. No. 5,115,202 describes a "spread-spectrum" approach toreducing noise in a chopper-stabilized op amp. The approach uses apseudo-random bit sequence ("PRBS") generator to pseudo-randomize thefrequency of the chopper switches. In single-chip integrated circuitapplications, however, this approach is not optimal. The PRBS generatorcreates digital noise on power supplies, which may interfere with thesignal being amplified by the primary op amp as well as affecting othercircuits on the chip. Moreover, the PRBS generator is onlypseudo-random, not truly random like the present invention. A PRBSgenerator has a limited capacity for outputting random numbers and willrepeat its output once that capacity is exhausted. A spectrum analysisof the output signal of a PRBS generator will yield some periodicsignal. Thus, it is not truly random. Finally, the degree of randomnessattained using the PRBS generator is not easily modifiable.

Accordingly, a need exists in integrated circuit applications for aclock with a true random, yet controllable, oscillation period. Use ofsuch a true random clock in, for example, a chopper-stabilized op amp tocontrol and randomize the amount of time between each switching of thechopper switches, would reduce the problem of interference due toaliasing of the noise created by the switching of the chopper switches.It would reduce such interference because the frequency of the switchingwould be random, not periodic, minimizing the generation of harmonics.Ideally, such a clock would not, in and of itself, create significant,if any, digital noise on power supplies.

SUMMARY OF THE INVENTION

In accordance with this invention, a random signal generator circuit isemployed in a chopper-stabilized operational amplifier on an integratedcircuit chip. The random signal generator outputs a voltage signalhaving a truly random voltage value within a selected range. A methodfor generating the truly random voltage signal is also disclosed. Therandom voltage signal is "truly" random in that its voltage value isunknown and unpredictable within the selected range, and repeats overtime only by random chance.

In one embodiment, the truly random voltage signal is a differentialvoltage signal that includes random white noise voltages, such as shotnoise voltages, in a selected frequency range. The random white noisevoltages are generated within the components of a first differentialtransconductance amplifier stage of the random signal generator circuit.This first amplifier stage is cascaded with two additional suchamplifiers. The first amplifier stage has no external input, and henceamplifies its own transistor shot noise or other is random white randomnoise voltages. The amplified white noise voltages are filtered by abandpass filter to select the frequency range of the truly randomvoltage signal. To prevent railing, the offset voltages of theamplifiers are nulled by feeding a voltage signal that is an integral ofthe output of the amplifiers back into the first amplifier.

In this embodiment, the random voltage signal generator circuit andmethod are used to form a random clock signal generator. The true randomvoltage signal that the above-mentioned random signal generator outputsis used to introduce a voltage control offset into an oscillator, suchas a rail-to-rail oscillator. Normally, the signal generated by such anoscillator would have a constant frequency, that is, it would have aconstant oscillation period or amount of time between oscillations andoutput voltage states. Inputting the true random voltage signal intosuch an oscillator causes the oscillation period of the oscillator tovary randomly. The amount of time between each alternating outputvoltage state of the oscillator, though truly random, is within aselected range of time because the value of the true random voltagesignal received by the oscillator is always within a selected range ofvalues. The alternating output voltage states of the oscillator form, orprovide a basis for forming, the series of truly random clockingsignals. An advantage of the embodiment of the random clock signalgenerator disclosed herein is that it generates little if any switchingnoise on DC power supplies.

As noted above, the random clock signal generator can advantageously beused in a chopper-stabilized operational amplifier. Chopper-stabilizedoperational amplifiers have one or more chopper switches in achopper-stabilizer circuit which is auxiliary to a primary operationalamplifier. Typically, such chopper switches repeatedly switch back andforth between two states or positions, with a constant amount of timebetween each switching, i.e., a constant switching frequency. Use of thetruly random clocking pulse to directly or indirectly control the amountof time between each switching of one or more of the chopper switchescauses the chopping period or frequency of the chopper switches also tobe truly random, but within a selected range. Truly randomly varying thechopping period or frequency of the switches reduces interference seenby the primary amplifier due to aliasing of noise created at thefrequency of the chopping. An advantage of the present invention is thatthe degree of randomness in the chopping, that is the range of theamount of variation in the chopping frequency or period is selectable,which allows one to optimize the performance of the chopper-stabilizedoperational amplifier.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a conventional chopper-stabilizedoperational amplifier.

FIG. 2 is a circuit diagram of a random signal generator circuit inaccordance with the invention.

FIG. 3 is a circuit diagram of a conventional rail-to-rail oscillatorcircuit.

FIG. 4 is a circuit diagram of a random clock signal generator inaccordance with the invention, including a random signal generator and aconventional rail-to-rail oscillator.

FIG. 5 is an alternative circuit diagram of a random signal generator inaccordance with the invention.

FIG. 6 is a circuit diagram of random clock signal generator inaccordance with the invention, including the alternative random signalgenerator and an alternative oscillator.

FIG. 7 is a circuit diagram of a chopper-stabilized operationalamplifier having a random clock signal generator.

FIG. 8 is a flow chart of steps in a method of controlling the amount oftime between switchings of a chopper switch in a chopper-stabilizedoperational amplifier.

DETAILED DESCRIPTION

FIG. 2 is a diagram of a random signal generator 20 in accordance withthe present invention. It includes operational amplifier 21, which has anoninverting ("+") input terminal 22, an inverting ("-") input terminal23, and an output terminal 24. Amplifier 21 is supplied with DC biasvoltages (not shown), but has no signal input. Capacitor C4 is connectedbetween noninverting terminal 22 and inverting terminal 21. A feedbackloop 25, which includes resistor R3 connected in parallel with capacitorC5, is connected between output terminal 24 and inverting ("-") inputterminal 23. Noninverting terminal 22 is grounded.

The capacitors and resistors of FIG. 2 provide a bandpass filter. Thebandpass filter regulates the range of frequencies of a voltage signalV_(RAN) output by random signal generator 20. Resistor R3 and capacitorC5 set the corner frequency of the low pass filter. Resistor R3 andcapacitor C4 set the corner frequency of the high pass filter.

Resistor R3 and capacitor C4 also function as an integration feedbackcircuit. They integrate the voltage signal output on line 24 byamplifier 21, and produce an integral voltage signal reflecting thatintegration step. The integral voltage signal is fed back via negativefeedback into inverting ("-") input terminal 23 of amplifier 21. Thisfeedback functions to null any offset voltages of amplifier 21.

Without an input signal, and with its offset voltages nulled, amplifier21 of random signal generator 20 amplifies random white noise voltageswhich are physically generated by electrical components, such astransistors or resistors within amplifier 21. These random white noisevoltages are generated, for example, by the application of DC biasvoltages to such components.

In one embodiment, a primary component of such random white noisevoltages is shot noise voltages generated by the DC biasing of bipolartransistors in a bipolar input stage (not shown) of amplifier 21. Shotnoise is a quantization error resulting from the fact that the number ofelectrons passing through a junction at different points in time varies.Another possible source of these random white noise voltages is thermalnoise due to resistors in series with the input stage. The random whitenoise voltages include random voltage signals in essentially allfrequencies. For a background discussion on such noise voltages, seeHorowitz, P. and Hill, W., "The Art of Electronics," pp. 286-307,Cambridge Univ. Press (New York 1980).

The output of random signal generator 20 is a truly random voltagesignal V_(RAN) that includes amplified random white noise voltages in aselected range of frequencies. Variables affecting the RMS voltage valueof V_(RAN) include the amount of noise generated, the amount ofamplification of the noise, and the corner frequencies of the bandpass.The amount of noise generated primarily depends on the amount of biascurrent selected to be passed through bipolar transistors in the inputstage of amplifier 21 and the selected value of any resistors selectedto be in series with the input stage. The amount of amplification ofthese random white noise voltages depends on the gain selected foramplifier 21. The value of the corner points of the bandpass depends onthe capacitance values selected for capacitors C4 and C5 of FIG. 2.

The RMS voltage value of the voltage signal V_(RAN) output by randomsignal generator 20 truly randomly varies within a selected range of RMSvoltage values. The RMS voltage value of V_(RAN) is unknown andunpredictable at any given point in time, i.e., is truly random, but itis always within the selected range. The range of possible values ofV_(RAN) is selected by the selections described in the precedingparagraph.

Random signal generator circuit 20 may be combined with a conventionalrail-to-rail oscillator to form a random clock signal generator. FIG. 3shows a conventional rail-to-rail oscillator. Conventional oscillator 30includes a differential op amp 31 connected to +5 and -5 volt suppliesand having an output terminal V_(out). A positive feedback loop 32consisting of equal-value resistors R4 and R5 is coupled between thenon-inverting terminal ("+") and output terminal of op amp 31. Anegative feedback loop 33 consisting of resistor R6 and capacitor C6 iscoupled between the inverting terminal ("-") and output terminal of opamp 31.

When power is initially provided to conventional oscillator 30, theoutput voltage V_(out) will be at either the +5 or -5 volt rail. For thediscussion that follows, we will assume that V_(out) is initially at the+5 volt rail. A toggle voltage of approximately 2.5 volts will appear atthe non-inverting terminal ("+") of op amp 31 via positive feedbackresistors R4 and R5. Current flowing through resistor R6 will chargecapacitor C6 towards the toggle voltage of 2.5 volts. As the invertingterminal ("-") of op amp 31 reaches 2.5 volts, V_(out) will swing low to-5 volts.

Once V_(out) toggles low to -5 volts, resistors R4 and R5 will pull thenon-inverting terminal ("+") of op amp 31 down to approximately -2.5volts. Capacitor C6 will then discharge and gradually pull the invertingterminal ("-") of op amp 31 to -2.5 volts. Just as the invertingterminal voltage reaches the voltage of the noninverting terminal, i.e.,-2.5 volts, op amp 31 toggles and V_(out) goes high to 5 volts. In thismanner, V_(out) oscillates rail-to-rail between 5 and -5 volts. The timebetween the rail-to-rail oscillations of oscillator 30 is constant,i.e., its period and frequency is constant, and is set by selection ofthe RC time constant of the negative feedback loop (i.e., the values ofresistor R6 and capacitor C6).

The oscillating rail-to-rail voltage outputs of oscillator 30 of FIG. 3may be used directly or indirectly to form a series of clock signals orpulses to directly or indirectly regulate the amount of time betweeneach of the successive switchings of the switches in achopper-stabilized operational amplifier, such as that shown in FIG. 1.In accordance with one embodiment the present invention, the frequencyor amount of time between the alternating rail-to-rail output voltagestates of such an oscillator may be made to truly randomly vary byinputting the truly random voltage signal V_(RAN) output by randomsignal generator 20 into oscillator 30.

FIG. 4 shows a random clock signal generator 40, formed of the randomsignal generator 20 of FIG. 2 and the conventional rail-to-railoscillator 30 of FIG. 3. (Components of FIGS. 2 and 3 which also appearin FIG. 4 are identified by the same numbers).

As described above with respect to FIG. 2, random signal generator 20outputs a truly randomly varying voltage signal V_(RAN), whose voltagevalue truly randomly varies within a selected range. In FIG. 4, output24 is connected to resistor R5, which in turn is connected to thenoninverting terminal ("+") of op amp 31. Inputting the true randomvoltage signal V_(RAN) into the noninverting terminal ("+") of op amp 31via resistor R5 truly randomly changes the toggle voltage of oscillator30. Accordingly, the amount of time necessary for capacitor C6 to reachthe toggle voltage also varies truly randomly. Hence, the amount of timebetween each of the successive rail to rail oscillations of oscillator30, and thus between each change of state of the clocking signal V_(out)output by random signal generator 40, will vary in a truly random mannerwithin a selected range of time. In other words, the rail-to-railvoltage signal V_(out) output by random clock signal generator 40appears to be truly randomly phase modulated, albeit in a controlledrange.

Although the exact transition time between each successive outputvoltage rail or state of random clock signal generator 40 is unknown andunpredictable, the range of the clock signal periods and the averageclock signal period will be known. The range of periods and the averageperiod are selected by the choice of the range of possible values forthe true random voltage signal V_(RAN), as is described above, and ofthe underlying oscillation period of oscillator 30, which is alsodescribed above.

FIG. 5 shows an alternative random signal generator 50 in accordancewith the present invention, which may be formed on a single integratedcircuit chip using a BICMOS fabrication process. It may, for example, beformed on the same chip as a differential oscillator to form a randomclock signal generator, and/or on the same chip as a chopper-stabilizedoperational amplifier whose switch closings are controlled by a seriesof clocking pulses output by the random clock signal generator.

Random signal generator 50 outputs a random differential voltage signal,V_(out),1 -V_(out),2, whose RMS voltage value at any point in time istruly random, i.e., unknown and unpredictable and repeating only byrandom chance, but is always within a selected range.

Random signal generator 50 includes three cascaded differentialtransconductance amplifier circuits or gain stages, which are denoted:G1, G2, and G3.

Amplifier G1 is a first or input stage, and includes a pair of bipolarnpn transistors Q1 and Q2. As is discussed below, the bases oftransistors Q1 and Q2 receive as an input via nodes F and G adifferential voltage signal reflecting the integral of the differentialvoltage output of the final noise amplifier stage, which in thisembodiment is the G3 stage. The differential output voltage of the G1stage is taken across the collectors of transistors Q1 and Q2.

Amplifier stage G2 is a second amplification stage, and includes a pairof bipolar npn transistors Q3 and Q4. Amplifier stage G3 is a thirdamplification stage, and includes a pair of bipolar npn transistors Q5and Q6. The bases of transistors Q3 and Q4 of the G2 stage receive as adifferential input the differential voltage output by the G1 stage. TheG2 stage outputs an amplified differential voltage signal reflecting itsinput. The output of the G2 stage is taken across the collectors oftransistors Q3 and Q4. The bases of transistors Q5 and Q6 of the G3stage receive as a differential input the differential voltage output bythe G2 stage. At the collectors of transistors Q5 and Q6, the G3 stageoutputs an amplified differential voltage signal reflecting the input tothe G3 stage. The true random differential voltage signal (V_(out),1-V_(out),2) output by random signal generator 50 is taken across thecollectors of transistors Q5 and Q6 of the G3 stage.

Returning to the G1 stage, pnp transistors Q7 and Q8, in combinationwith resistors R7 and R8, source a DC bias current to the collectors oftransistors Q1 and Q2. Resistors R7 and R8 are connected across thebase/collector junctions of transistors Q7 and Q8, respectively. Thisarrangement is suited for low voltage operation, since transistors Q7and Q8 function to ensure that the average DC bias voltage at thecollectors of Q1 and Q2 is constant at one diode drop below the DC biassupply voltage.

The G2 stage and the G3 stage have identical source current arrangementsat the collectors of their respective transistor pairs. Bias current forthe G2 stage is sourced by pnp transistors Q9 and Q10 and resistors R9and R10, and bias current for the G3 stage is sourced by pnp transistorsQ11 and Q12 and resistors R11 and R12.

Current sources I₁, I₂, and I₃ bias the G1, G2, and G3 stages. Each ofthese current sources is generated by a band gap generator, whichensures that the G1, G2, and G3 stages have a constant gain over processand temperature. The amount of gain of the G1, G2, and G3 stages isselected by the amount of current sourced by I₁, I₂ and I₃,respectively.

Capacitor C7 across the collectors of transistors Q5 and Q6 of the G3stage forms part of the bandpass filter of random signal generator 50.As an example, a capacitor having a value of approximately onepico-Farad may be used.

The embodiment of FIG. 5 includes a differential feedback loop whichincludes an integrator 52 and conductive lines 51 and 54. Integrator 52is shown within the dashed box in FIG. 5. The true random differentialvoltage signal output at the collectors of transistors Q5 and Q6 of theG3 stage is input to integrator 52 at the bases of transistors Q13 andQ14 of the G4 stage and is integrated by integrator 52.

Integrator 52 of FIG. 5 includes a differential transconductanceamplifier stage, denoted G4, and a floating capacitor circuit 53. The G4stage includes a pair of bipolar npn transistors Q13 and Q14. G4 isbiased by current source I₄, which is a band gap generated currentsource. The G4 stage has a very low gain and functions like a very largeresistor.

The bases of transistors Q13 and Q14 are connected to the collectors oftransistors Q5 and Q6 of the G3 stage, and thus receive as an input thetrue random differential voltage signal output by the G3 stage. Thecollectors of transistors Q13 and Q14 of the G4 stage are connected tothe collectors of pnp transistors Q15 and Q16, respectively, of floatingcapacitor circuit 53. The cooperation and functioning of the G4 stageand floating capacitor circuit 53 of integrator 52 to produce adifferential voltage signal (V_(i),1 -V_(i),2) reflective of theintegral of the output of the G3 stage is addressed below.

The differential integral voltage signal (V_(i),1 -V_(i),2) output byintegrator 52 at the drains of P channel transistors MP5 and MP6 is fedas a sole input back into the bases of transistors Q1 and Q2 of the G1stage via lines 51 and 54 and nodes F and G. The base of transistor Q1is electrically connected to node F, and the base of transistor Q2 iselectrically connected to node G. Nodes F and G are also connected toground through resistors R13 and R14, respectively, diode D1, andresistor R15. Example values of resistors R13 and R14 are 1 k-ohm.

Feeding the differential integral of the G3 output back into the G1stage, specifically to the bases of transistors Q1 and Q2, practicallynulls the offsets of the G1, G2, and G3 stages. This prevents railing ofthe output of random signal generator 50 due to the undesired, butlargely unavoidable, offset voltages of the amplifier stages.

With the offset voltages of the G1, G2, and G3 amplifier stages nulledin such a manner, and without any other input to the bases oftransistors Q1 and Q2 other than from nodes F and G, the G1 stageamplifies and outputs truly random white noise voltages that inherentlyresult from the application of current to circuit components. In theembodiment of FIG. 5, these random white noise voltages include shotnoise voltages generated by the application of DC bias current totransistors Q1 and Q2 of the G1 stage, together with any thermal noisevoltages of resistors R13 and R14. The amount of noise so generated isselected by the amount of bias current and the value of resistors R13and R14. Other on-chip sources of random white noise voltages could betapped in alternative embodiments.

The truly random white noise voltage signals amplified and output by theG1 amplifier are subsequently amplified by the G2 and G3 stages.Components within the G2 and G3 stages also generate such random whitenoise voltages, but their random white noise voltages are drowned withinthe amplified output of the G1 stage.

Random signal generator 50 also includes a bandpass filter, which setsthe frequency range or bandwidth of the differential voltage signal(V_(out),1 -V_(out),2) that random signal generator 50 outputs.

As discussed above, G1 amplifies its own random white noise voltages,which essentially includes all frequencies, and the G2 and G3 stagessuccessively amplify the differential voltages output by the G1 stage.The bandpass filter filters out all but a selected frequency range ofthese random white noise voltages. In other words, the random whitenoise voltage signal output by random signal generator 50 reflects onlya selected range of the white noise frequencies originally output by theG1 stage. The high frequency corner point of the bandpass is set byresistors R11 and R12 working into capacitor C7, and is selected by theresistance or capacitance values chosen for those components. The lowfrequency corner point of the bandpass is set by the effective impedanceof transistors Q13 and Q14 of the G4 stage working into capacitor C8,and is selected by the effective impedance or capacitance values chosenfor those components.

The ultimate output of random signal generator 50, taken at thecollectors of transistors Q5 and Q6 of the G3 stage after theabove-described steps of noise voltage generation, amplification, bandpass filtering, and integration feedback, is a true random differentialvoltage signal (V_(out),1 -V_(out),2) comprised of selectively amplifiedrandom white noise voltages in a selected range of frequencies.V_(out),1 -V_(out),2 is a truly random voltage signal because it iscomprised of amplified truly random white noise voltages (e.g., shotnoise voltages) in a selected range of frequencies. The RMS voltagevalue of this differential output voltage signal truly randomly varieswithin a selected range. It is unknown, unpredictable, and repeats overtime only by random chance.

The RMS voltage value of differential voltage signal V_(out),1-V_(out),2 is always within a selected range. In other words, the degreeof randomness is controlled. The range of RMS voltage values of thisrandom differential voltage signal is selected by the choice ofcomponents and/or voltage or current values within random signalgenerator 50, as discussed above.

Example design choices for random signal generator 50 of FIG. 5 include:a collector current through each of transistors Q1, Q2, Q3, Q4, Q5, andQ6 of approximately 3 micro-amps; resistance values for resistors R7,R8, R9, and R10 of approximately 100 k-ohms, and for resistors R11 andR12 of approximately 200 k-ohms; a capacitance value for capacitor C7 ofapproximately 1 pico-Farad, and for capacitor C8 of approximately 6pico-Farads; and, a collector current through each of transistors Q13and Q14 of approximately 183 pico-amps. An example average value forV_(out),1 -V_(out),2 is 10 milli-volts RMS, and an example range ofpossible values for V_(out),1 -V_(out),2 is between about 5 and 20milli-volts RMS.

Integrator 52, i.e., the G4 stage and floating capacitor circuit 53, isin accordance with the circuit shown in FIG. 2 of U.S. patentapplication Ser. No. 08/527,401, entitled "Floating CapacitorDifferential Integrator," now U.S. Pat. No. 5,793,242 which applicationis incorporated herein by reference. The reader should refer to thatapplication for a detailed description of integrator 52.

Referring to integrator 52 of FIG. 5 of the present application, thecollectors of transistors Q13 and Q14 of the G4 stage are connected tothe collectors of pnp transistors Q15 and Q16, respectively, of floatingcapacitor circuit 53. Transistors Q13 and Q14 sink current fromtransistors Q15 and Q16, respectively, responsive to the differentialvoltage applied to the bases of transistors Q13 and Q14. Transistors Q15and Q16 provide nodes H and J, respectively, with a common mode currentvia P-channel pass transistors MP1 and MP2, respectively. Currentsources I₅ and I₆ supply pass transistors MP1 and MP2, respectively.Current sources I₇ and I₈, supply P-channel transistors MP3 and MP4,respectively. MP3 and MP4 control the conductive states of transistorsMP1 and MP2, respectively. P-channel transistors MP5 and MP6, along withresistors R16 and R17, form a common mode feedback loop. The drains oftransistors MP5 and MP6 are connected to nodes F and G, respectively, asdiscussed above. Resistor R18 sources current to transistors MP5 and MP6via resistors R16 and R17, respectively. An integrating capacitor C8 isconnected between nodes H and J. Capacitor C8 is fabricated so as tohave equal parasitic leakages on each of its plates, as modeled bydiodes D2 and D3. Capacitor C8 may be replaced by two capacitorsconnected in parallel.

Integrator 52 is fully differential. It responds only to differentialcurrents and voltages, and ignores common mode currents and voltages.Its symmetrical design allows it to balance increases in voltages on oneside of the circuit, e.g., the side of node H, with equal decreases involtages and/or currents on the other side of the circuit, e.g. the sideof node J. Since the common mode or average voltage of nodes H and Jremains constant, capacitor C8 "floats" at a constant voltage.

Referring again to FIG. 5 of the present application, when thedifferential voltage output by the G3 stage is input to the G4 stage,i.e., input to the bases of transistors Q13 and Q14, integrator 52produces a differential voltage output signal (V_(i),1 -V_(i),2), whichappears across the drains of transistors MP5 and MP6. This differentialintegral voltage signal is provided via lines 51 and 54 to nodes F andG, and in turn to the bases of transistors Q1 and Q2 of the G1 stage.

Integrator 52 achieves a relatively high RC time constant, whileminimizing the size of capacitor C8, because integrator 52 maintains thecollector voltages of transistors Q15 and Q16 at a substantiallyconstant value. Since the collector voltages of transistors Q15 and Q16are insensitive to changes in the respective drain voltages oftransistors MP1 and MP2, the collectors of transistors Q15 and Q16 arebuffered to act as nearly ideal current sources. That is, the drains oftransistors MP1 and MP2 exhibit impedances approaching infinite. Itfollows, then, that the effective impedance across capacitor C8, as seenlooking in from nodes H and J, approaches infinite. The above describedstructure realizes impedances across capacitor C8 on the order ofterra-ohms or higher.

The arrangement of random signal generator 50 may be varied. Forexample, the number of differential transconductance amplifier stagesmay be one, two, three (as shown), or more than three. As anotherexample, the design of integrator 52 may be changed. A conventionalresistor and capacitor may be used as an integrator circuit.

FIG. 6 of the present application shows a random clock signal generator60 comprised of a random signal generator block 50, an interface circuitincluding a differential amplifier stage 61, and a differentialoscillator 62. The random signal generator block 50 of FIG. 6 representsthe circuit of random signal generator 50 of FIG. 5 of the presentapplication. Random signal generator 50 outputs a true randomdifferential voltage signal V_(OUT),1 -V_(OUT),2, which includesamplified random white noise voltages in a selected range offrequencies.

Returning to FIG. 6, between random signal generator 50 and oscillator62 is an interface circuit, comprised of differential amplifier stage61, which includes pnp transistors Q15 and Q16. Current source I₉provides bias current for transistors Q15 and Q16. I₉ is a band-gapgenerated current source. An example collector current for each oftransistors Q15 and Q16 is 2.1 micro-amps.

The bases of transistors Q15 and Q16 receive the true randomdifferential voltage signal V_(OUT),1 -V_(OUT),2 output by random signalgenerator 50. The collectors of transistors Q15 and Q16 are connectedvia lines 65 and 66 to the emitters of transistors Q17 and Q18, at nodesM and N, respectively. Identical degeneration resistors R19 and R20 areconnected between nodes M and N, respectively, and ground. An examplevalue for each of resistors R19 and R20 is 100 k-ohms. An exampleaverage voltage drop across each of resistors R19 and R20 isapproximately 200 milli-volts.

Differential amplifier 61 effects a DC level shift between random signalgenerator 50, whose output has a DC component of about one volt, anddifferential oscillator 62. The emitter currents of transistors Q17 andQ18 of oscillator 62 have a DC component of approximately zero.

Oscillator 62 of FIG. 6 is the same design as the oscillator shown inFIG. 2 of application Ser. No. 08/527,400, which application isincorporated herein by reference, now U.S. Pat. No. 5,600,283, exceptfor degeneration resistors R19 and R20. Resistors R19 and R20 areidentically sized, and are connected between the collectors oftransistors Q17 and Q18, respectively, and ground. The reader shouldrefer to incorporated application Ser. No. 08/527,400 now U.S. Pat. No.5,600,283 for a detailed description of oscillator 62.

Within oscillator 62, NPN transistors Q17, Q18, Q19, Q20, Q21, and Q22form a gain stage which is connected between the right and left outputterminals, V_(OUT),R and V_(OUT),L, respectively, of oscillator 62. NPNtransistors Q23 and Q24, together with resistors R_(TH),L and R_(TH),R,provide a differential positive feedback. PNP transistors Q25 and Q26form a differential comparator, while capacitor C9 and a differentialtransconductance amplifier or "gm" stage formed by NPN transistors Q27and Q28 provide a differential negative feedback. The gm stage is drivenby current source I_(gm) which, in turn, determines the slew rate ofcapacitor C9. Current source I_(TH) and resistors R_(TH),R and R_(TH),Ldetermine the toggle voltage of capacitor C9, while resistors R21 andR22 determine the common mode, or average, voltage of capacitor C9.P-channel pass transistors MP7 and MP8 buffer capacitor C9. Currentsource I_(b) biases transistors Q25 and Q26. PNP transistors Q29 andQ30, along with biasing resistor R23, form two equal current sources toprovide current to both nodes of capacitor C9. Oscillator 62 provides adifferential output voltage at output terminals V_(OUT),R and V_(OUT),L.

The output voltages at output terminals V_(OUT),R and V_(OUT),L arealternatively in high and low states. There is an amount of time betweeneach output state. In a first state, V_(OUT),R is high and V_(OUT),L islow. In a second state, V_(OUT),R is low and V_(OUT),L is high. WhenV_(OUT),R is high, its voltage value is approximately two diode dropsabove ground (V_(be) of transistor Q18+V_(be) of Q22), and V_(OUT),L isonly one diode drop above ground (i.e., V_(OUT),R -V_(be) of Q20). WhenV_(OUT),L is high, its voltage value is approximately two diode dropsabove ground (V_(be) of Q17+V_(be) of Q19), and V_(OUT),R is only onediode drop above ground (i.e., V_(OUT),L -V_(be) of Q21).

Assume for a moment that there is no connection between amplifier 61 andoscillator 62 and no resistors R19 and R20. As described in incorporatedapplication Ser. No. 08/527,400 now U.S. Pat. No. 5,600,283, oscillator62 normally oscillates between its output states when the base voltagesof transistors Q25 and Q26, which form a comparator, are approximatelyequal. There are two inputs to the comparator formed by transistors Q25and Q26, which cause oscillator 62 to toggle. First, there is a positivefeedback through transistors Q23 and Q24 working into resistors R_(TH),Land R_(TH),R. Second, there is a negative feedback through transistorsQ27 and Q28 working into capacitor C9, which is buffered by p-channeldevices MP7 and MP8. The collector currents of transistors Q17 and Q18are equal, and the amount of time between the alternating output statesof oscillator 62 is always the same amount of time, i.e., theoscillation period and frequency are constant. The oscillator's periodand frequency are selected by component choices, as is described inincorporated application Ser. No. 08/527,400 now U.S. Pat. No.5,600,283.

In the embodiment of FIG. 6 of the present application, a voltagecontrolled offset is introduced into oscillator 62 at nodes M and N.Recall that random signal generator 50 of outputs a true randomdifferential voltage signal having a truly random RMS voltage valuewithin a selected range. The introduction of this truly random varyingdifferential voltage signal from random signal generator 50 viaamplifier 61 into oscillator 62 at nodes M and N upsets the matchingthat would otherwise exist between the collector currents of transistorsQ17 and Q18. The collector currents of transistors Q17 and Q18 willtruly randomly vary because of the truly randomly varying voltage dropsacross resistors R19 and R20. This in turn affects the base voltagetoggling of transistors Q25 and Q26. It is as if a truly randomlyvarying voltage source is placed in series with the inputs of thecomparator formed by transistors Q25 and Q26. Accordingly, the amount oftime it takes until the base voltages of transistors Q25 and Q26 becomeequal, causing oscillator 62 to toggle between its output states, willalso truly randomly vary. Hence, the amount of time between successiveoscillations of oscillator 62 and between its alternating output voltagestates will truly randomly vary within a selected range of time oneither side of the average oscillation period of oscillator 62, and theamount of time between each oscillation will repeat only by randomchance. The amount of time variation between each oscillation andalternating output state is within a controlled range of time, forexample, ±100 microseconds around the average period, because the RMSvoltage value of the true random voltage signal output by random signalgenerator 50 is always within a selected range.

The range of time variation in the oscillation periods may be adjustedby changing the true random voltage signal V_(out),1 -V_(out),2 outputby random signal generator 50. For example, the amount amplification ofthe random white noise voltages or the range of frequencies passed bythe bandpass filter of random signal generator 50 may be changed.

As an example, random clock signal generator 60 may have an averageoscillation period of 250 microseconds, with a truly random variationabout the average in a range of plus or minus approximately 100micro-seconds. In other words, the oscillator period truly randomlyvaries within a range of time of between 150 and 350 microseconds.

The oscillating first and second voltage output states of random clocksignal generator 60 may be used, directly or indirectly, to form aseries of clocking pulses to regulate the timing of successiveswitchings of the chopper switches in a chopper-stabilized operationalamplifier, among other applications. Using a random clock signalgenerator circuit having such a random oscillation period avoids orreduces the problem of noise caused by a periodic switching of thechopper switches. It also avoids the DC noise that would be created bythe use of a PRBS generator, because random clock signal generator 60,including random signal generator 50 and oscillator 62, draws only low,constant DC currents, for example, 16 micro-amps. Moreover, unlike thepseudo-random clocking pulses obtainable using a PRBS generator, theamount of time between each of the oscillating first and second outputvoltage states of random clock signal generator 60 is truly random andrepeats only by random chance. The degree of randomness is alsoselectable, as described above.

FIG. 7 shows an embodiment of a chopper-stabilized operational amplifier70 in accordance with the present invention. FIG. 7 includes thechopper-stabilized amplifier of FIG. 1, except that clock 15 of FIG. 1is replaced by a random clock signal generator 60. The switches ofchopper-stabilized operational amplifier 70 may be low noise chopperswitches, such as are described in above-referenced application Ser. No.08/811,063, now U.S. Pat. No. 5,959,498 issued on Sep. 28, 1999.

Random clock signal generator 60 of FIG. 7 includes a random signalgenerator 50, an interface amplifier 61, and an oscillator 50, whichrepresent the analogous circuits of FIGS. 5 and 6. The random signalgenerator 50 of FIG. 7 outputs a true random differential voltage signalincluding amplified random white noise voltages in a selected range offrequencies. The instantaneous RMS voltage value of this signal is trulyrandom, and repeats only by random chance. It is, however, always withina selected range.

Oscillator 62 of FIG. 7 oscillates between and outputs a first outputvoltage state and a second output voltage state. The time intervalsseparating each of the transitions back and forth between the first andsecond output voltage states are truly random in duration, but liewithin a selected range of times. These alternating output voltagestates may be used, directly or indirectly, as a clocking pulse tosignal the times at which the chopper switch or switches of FIG. 7should switch between states or positions.

In another aspect, the present invention includes methods of controllingthe amount of time between each switching of one or more switches in achopper-stabilized operational amplifier. FIG. 8 is a flow chart of oneembodiment such methods. A first step includes selecting a range of timewithin which the switch (or switches in an alternative embodiment)should switch back and forth between a first state or position and asecond state or position. For example, in an embodiment such as in FIG.7, the first state of switch 12 is node A, and the second state ofswitch 12 is node B. The amount of time between a switching from thefirst state to the second state, or vice versa, may be, for example,anywhere between 150 microseconds and 350 microseconds, with an averagetime between each switching of 250 micro-seconds. A second step is torepeatedly generate a clocking signal to indicate, directly orindirectly, when the switch should switch between states.

In accordance with the present invention, a third step is to trulyrandomly vary, within the selected range of time, the amount of timebetween each clocking signal, and hence between each successiveswitching of the chopper switch.

In greater detail, the above method may include steps of creating randomwhite noise voltages, such as shot noise voltages, on the sameintegrated circuit chip on which the chopper-stabilized operationalamplifier is implemented. For example, shot noise voltages may begenerated by DC biasing the bipolar transistors in a differentialtransconductance amplifier, such as bipolar transistors Q1 and Q2 in theG1 stage of FIG. 5 of the present application. The amount of shot noisevoltages so generated may be controlled by selecting the amount of biascurrent. Steps of amplifying and bandpass filtering these random whitenoise voltages are then performed. The amount of amplification may beselected by choosing an amplifier having an appropriate gain. Thebandpass filtering step, which may be performed with an RC network orequivalent, ensures that a selected frequency range of the random whitenoise voltages is passed. Random white noise voltages at frequenciesoutside the selected frequency range are filtered out. The net result isa true random voltage signal, which may be a differential voltagesignal, comprised of amplified random white noise voltages in a selectedrange of frequencies. This voltage signal, which will be referred to asa true random voltage signal, has a RMS voltage value which has a trulyrandom value within a selected range at any point in time. The RMSvoltage value of this signal repeats over time only by random chance. Anexample average value of this true random voltage signal, which may begenerated by employing random signal generator 50 of FIG. 5, is about 10milli-volts RMS, and truly randomly ranges in value from about 5 to 20milli-volts RMS.

The true random voltage signal so generated may be input, in a nextstep, to an oscillator to form a series of true random clocking signalsor pulses to directly or indirectly control when the chopper switchesshould successively switch from one state to the other state. Theoscillator provided should be capable of oscillating between andoutputting a first output voltage state and a second output voltagestate, with the alternating output voltage states serving as theclocking signals. As an example, oscillator 30 of FIG. 3 or oscillator62 of FIG. 6 may be provided.

A next step is to truly randomly vary the amount of time between eachsuccessive oscillation and output voltage state of the oscillator sothat the amount of time truly randomly varies within a selected range oftime. A circuit such as the random clock signal generators of FIGS. 4and 6 may be employed to implement this method. Referring to FIG. 4,oscillator 30 has, in a case where no input signal is provided fromrandom signal generator 20, a constant standard oscillation period whichis selected by the RC time constant of the negative feedback loop ofoscillator 30. Inputting a truly randomly varying voltage signal fromrandom signal generator 20 into oscillator 30 at resistor R5 causes theamount of time between oscillations of oscillator 30 to truly randomlyvary, within a range of time, which range reflects the range of possiblevalues of the true random voltage signal. The range of possible valuesof the true random voltage signal is established by the previouslymentioned steps of noise voltage creation, amplification, and bandpassfiltering. The true random clocking signal is formed, directly orindirectly, from the alternating output voltage states output by theoscillator.

In an embodiment such as is shown in FIG. 5, there are additional stepsof providing three cascaded differential transconductance amplifiers onthe same integrated circuit chip as the chopper-stabilized operationalamplifier. Steps of creating random white noise (e.g., shot noise)voltages within a first amplifier stage, and then amplifying and bandpass filtering such noise voltages are then performed, so that a truerandomly varying differential voltage signal is formed which has a trulyrandom RMS voltage value within a selected range of RMS voltage values.This signal may then be input into an oscillator, as described above, toform and output a series of clocking signals, each of which has a trulyrandom period within a selected range of periods.

An additional step of nulling the offsets of the amplifier stages may beundertaken to prevent railing of the amplifier outputs. For example, inthe embodiment of FIG. 5, this nulling step is performed by feeding anintegral of the output voltage of amplifiers G1-G3 back into the firstamplifier stage, specifically to the bases of transistors Q1 and Q2 ofthe G1 stage.

The embodiments described above are but examples of the presentinvention. Artisans will recognize that variations are possible.

I claim:
 1. A chopper-stabilized operational amplifier implemented on asingle integrated circuit chip comprising:an operational amplifier onsaid chip; a chopper-stabilizer circuit on said chip auxiliary to theoperational amplifier, wherein the chopper-stabilizer circuit includes achopper switch that oscillates between a first state and a second state,with a first amount of time between each oscillation; and a clock signalgenerator circuit on said chip, wherein the clock signal generatorcircuit includes a first component that generates white noise voltagesupon application of a DC bias current, and the clock signal generatorcircuit outputs a first voltage signal derived from the white noisevoltages; and wherein the first voltage signal is provided directly orindirectly to the chopper stabilizer circuit, and the first amount oftime between oscillations of the chopper switch varies randomly within aselected range based at least in part on the first voltage signal. 2.The chopper-stabilized operational amplifier of claim 1, wherein theclock signal generator circuit includes a first amplifier and a bandpassfilter, wherein said first amplifier amplifies said white noise voltagesand the bandpass filter filters said white noise voltages.
 3. Thechopper-stabilized operational amplifier of claim 2, wherein the firstcomponent is external to said first amplifier.
 4. The chopper stabilizedoperational amplifier of claim 2, where the first component is internalto said first amplifier.
 5. The chopper-stabilized operational amplifierof claim 2, wherein the clock signal generator circuit includes a secondamplifier connected between said first amplifier and said bandpassfilter, wherein the second amplifier receives amplified white noisevoltages output from the first amplifier, and the bandpass filterreceives an output of the second amplifier.
 6. The chopper-stabilizedoperational amplifier of claim 1, wherein said first voltage signaloscillates between a first voltage state and a second voltage state withan amount of time between each oscillation, wherein the amount of timevaries randomly within a selected range.
 7. The chopper-stabilizedoperational amplifier of claim 2, wherein the clock signal generatorcircuit includes an oscillator;wherein said oscillator receives as aninput a second voltage signal derived at least in part from the whitenoise voltages amplified by the first amplifier and filtered by thebandpass filter; said oscillator produces the first voltage signal basedat least in part on the second voltage signal; and the first voltagesignal oscillates between a first voltage state and a second voltagestate with an amount of time between each oscillation, wherein theamount of time varies randomly within a selected range.
 8. Thechopper-stabilized operational amplifier of claim 1, wherein the clocksignal generator circuit includes a random signal generator circuit;saidrandom signal generator circuit including a first amplifier and abandpass filter; wherein said first amplifier amplifies said white noisevoltages and outputs a second voltage signal; and the bandpass filterreceives as an input the second voltage signal or a signal derived fromthe second voltage signal, filters said input, and outputs a thirdvoltage signal.
 9. The chopper-stabilized operational amplifier of claim8, further comprising an integrator circuit on said chip, wherein theintegrator circuit receives as an input the third voltage signal or asignal derived from said third voltage signal, integrates said input,and outputs an integrated fourth voltage signal to the first amplifier.10. The chopper-stabilized operational amplifier of claim 9, wherein therandom clock signal generator circuit includes an oscillator;theoscillator receives as an input the third voltage signal or a signalderived from the third voltage signal; and the oscillator outputs thefirst voltage signal based at least in part on the third voltage signal;and the first voltage signal oscillates between a first voltage stateand a second voltage state with an amount of time between eachoscillation, wherein the amount of time varies randomly within aselected range.
 11. The chopper-stabilized operational amplifier ofclaim 1, wherein the clock generator circuit draws only constant DC biascurrents from a power supply.
 12. The chopper-stabilized operationalamplifier of claim 2, wherein the clock generator circuit draws onlyconstant DC bias currents from a power supply.
 13. A method performed ona single integrated circuit chip of timing the switchings of a chopperswitch in a chopper-stabilized operational amplifier implemented on thesingle integrated circuit chip comprising:creating white noise voltageson said integrated circuit chip; amplifying and bandpass filtering saidwhite noise voltages on said chip to produce a first voltage signal onsaid chip, said first voltage signal having a randomly varying voltagevalue within a selected range of voltage values; controlling theswitching of the chopper switch on said chip with the first voltagesignal or a signal derived from the first voltage signal; and switchingthe chopper switch between a first state and a second state with arandomly varying amount of time between each switching, wherein theamount of time varies randomly within a selected range.
 14. The methodof claim 13, further includingproviding an oscillator on said integratedcircuit chip; inputting said the amplified and bandpass filtered whitenoise voltages, or a signal derived therefrom, into said oscillator; andproducing said first voltage signal with said oscillator, wherein saidfirst voltage signal oscillates between a first voltage state and asecond voltage state with a randomly varying amount of time between eachoscillation, said amount of time varying randomly within a selectedrange of time.
 15. A randomized clocking circuit implemented on a singleintegrated circuit chip for generating a randomized clocking signal thatis provided to a second circuit on the same chip comprising:a componenton said chip that generates white noise voltages upon application of aDC current; an amplifier on said chip that amplifies said white noisevoltages; a bandpass filter on said chip that filters said amplifiedwhite noise voltages; an oscillator on said chip that receives theamplified and bandpass filtered white noise voltages, or a signalderived therefrom, and outputs said randomized clocking signal, whereinsaid randomized clocking signal oscillates between a first voltage stateand a second voltage with a random amount of time between oscillations,said amount of time varying randomly within a selected range; and asecond circuit on said chip that receives and utilizes said randomizedclocking signal.
 16. The randomized clocking circuit of claim 15,wherein said randomized clocking circuit draws only constant DC biascurrents from a power supply.
 17. The randomized clocking circuit ofclaim 15, wherein the second circuit is a control for a chopper switchof a chopper-stabilized operational amplifier.
 18. The circuit of claim15, further comprising an integrator circuit on said chip, wherein saidintegrator circuit is in a feedback connection to the amplifier.
 19. Thecircuit of claim 18, wherein the second circuit is a control for achopper switch of a chopper-stabilized operational amplifier.
 20. Amethod of producing and utilizing on a single integrated circuit chip arandomized clocking signal that oscillates between a first voltage stateand a second voltage state with a randomly varying amount of timebetween each oscillation comprising:creating white noise voltages onsaid integrated circuit chip; amplifying and bandpass filtering saidwhite noise voltages to produce a first voltage signal, said firstvoltage signal having a randomly varying voltage value within a selectedrange of voltage values; providing an oscillator on said integratedcircuit chip, said oscillator capable of oscillating between a firststate and second state; inputting the first voltage signal or a signalderived therefrom into said oscillator; and controlling the oscillationsof the oscillator with said first voltage signal or the signal derivedtherefrom, so that the oscillator oscillates between the first state andthe second state with a randomly varying amount of time between eachoscillation, wherein said amount of time varies randomly within aselected range; outputting said randomized clocking signal from saidoscillator based on whether the oscillator is in the first state or thesecond state; and providing said randomized clocking signal to a secondcircuit on the same chip, and utilizing said randomized clocking signalwithin said second circuit.
 21. The method of claim 20, wherein saidrandomized clocking signal is produced entirely within circuits on saidchip, and further comprising providing only constant DC bias currents toall circuits used to produce said randomized clocking signal.
 22. Themethod of claim 20, further comprising providing an amplifier on saidchip, and integrating and feeding back said first voltage signal or asignal derived therefrom to said amplifier.
 23. A circuit implemented ona single integrated circuit chip for generating a randomized voltagesignal having a random voltage value within a selected rangecomprising:a component on said chip that generates white noise voltagesupon application of a DC current; a first amplifier on said chip thatamplifies said white noise voltages; a bandpass filter on said chip thatfilters said amplified white noise voltages; an integrator circuit onsaid chip in a feedback connection with said amplifier; and wherein saidcircuit outputs an amplified, bandpass filtered voltage signal having arandom voltage value that varies randomly within a selected range. 24.The circuit of claim 23, further comprising a second amplifier on saidchip between said first amplifier and said bandpass filter.
 25. Thecircuit of claim 24, further comprising a third amplifier on said chipbetween said second amplifier and said bandpass filter, and wherein saidintegrator circuit nulls any offset voltages of said second and thirdamplifiers.
 26. The circuit of claim 23, wherein said circuit draws onlyconstant DC bias currents from a power supply.